Softrock Lite 6.2
Adventures in Electronics and Radio
Elecraft K2 and K3 Transceivers
July 2007 Updates
28 July 2007
- Crispino, I5XWW, has supplied me with some Bourns
LM-NP-1001-B1 600Ω : 600Ω transformers to evaluate for use with a
Softrock.(Data sheet at
http://www.bourns.com/PDFs/LMNPLP.pdf) I've provided extensive
measurements at my Softrock page, but my
conclusion is that these transformers will work if you have the correct
terminating impedance�not 600 ohms, but something closer to 10K. Their high
sensitivity to terminating impedance, however, requires considerable
experimentation with your particular sound card before providing 96 KHz
bandwidth. And, depending on your sound card's input impedance, they may not
be acceptable at all. Hence, I cannot recommend these transformers as "plug
and play" for maximum bandwidth. For 24 KHz operation, almost any reasonable
impedance is satisfactory, however.
- I've mentioned on my
lattice crystal filter page that
anyone more than casually interested in filter design should have a copy of
Handbook of Filter Synthesis by Anatol I. Zverev in their library, but
lamented the fact that it was nearly a $300 book. Brad, AL4T, wrote to say
that Zverev is now available in a lower cost paperback edition. Indeed, Amazon
carries it for $58, which is only about twice what I paid for the hardbound
edition in 1975. To repeat�this book belongs in your personal library if
you are interested in filter design. If you are not an engineer,
don't be put off by all the equations. There are plenty of tables and
discussions you can use to design filters without delving deeply into the
mathematics. But, if you take the time and wade through the math, you will
have an even better understanding of filter design. Thanks again to Brad for
alerting me to the lower cost edition.
- Graham, KE9H, also writes about my island pad
capacitance measurements. I had assumed (always dangerous) that the thin PCB
stock I acquired surplus had 1 oz. copper foil. Graham says it's almost
certainly 0.5 oz foil in its present form, as the additional copper for
a 1 oz delivered finish is supplied during the electroplating process (traces
and holes) as the raw stock is made into a completed PCB. Hence, instead of
1.4 mils copper thickness, it's 0.7 mils. (0.036 mm and 0.018 mm,
respectively.) This changes my pad capacitance calculations slightly, as the
fiberglass portion of the sandwich increases. I've revised the calculations
accordingly on my Prototype page.
27 July 2007
Geoff, GM4ESD, has asked how much capacitance exists
between an island pad and ground, assuming the pad is cut on a piece of
double-sided PCB stock and that the top and bottom copper planes are tied
together, as is normally the case. The same question, of course, applies
to conventional Manhattan-style construction with glued-on bits of PCB stock on
top of a copper PCB surface.
I've added a section to my
Prototype page presenting both theoretical and measured values for island
26 July 2007
Because my shop has spread over most of the basement, it's
common to move a particular piece of test gear from one end to the other. I've
grown fond of scope carts (and own two), but several pieces of gear I own are
too large for the carts I see on the surplus market.
There's a reasonably priced instrument cart available from
Easy-Up (web site brochure at
I now have five of these, two flat top and three slope top "instrument carts,"
acquired over the years. The dealer I've used is Electronix Express, and the
carts are shown at
http://www.elexp.com/am_easy.htm. These carts are moderately
reasonably priced, in the $71 to $85 range, depending on size and finish. I've
found some on-line outlets selling these carts at $20 or more higher price, so
shop carefully, if you decide to purchase one.
The carts are available in either unfinished hard
board or gray plastic laminate over particle board, with the laminate finish
being a few dollars more expensive. All mine are plastic laminate finish. The
carts have 3" casters for mobility. They come knocked down, and it takes about
20 to 30 minutes to assemble one. Normal hand tools are necessary.
Until recently, I've found the smaller size sloped cart
adequate. However, the HP3562A Dynamic Signal Analyzer I recently acquired
requires the large EZ45-8G cart, 23" wide x 19" deep. However, the 3562's feet
are more than 19" spaced, so I added a 1/2" thick 23" x 24" secondary top to the
EZ45-8G cart. I also added a blocking strip at the rear of the top to prevent
the 3562A from sliding backward off the cart. Normally, the Easy-Up carts
retain instruments with a rear stop bar, made from steel tubing, with rubber
pads. However, the 3562A is so large and heavy that the stop bar is not
adequate, particularly after I relocated it to the rear of the cart, made
necessary by the shelf extension.
The quality of these carts is not equal to a purpose-made
Tektronix or HP oscilloscope cart, of course, but they are serviceable,
particularly in the basement shop and the price is reasonable. The 3562A weighs
70 pounds or so and the cart is quite stable with it in place.
HP3562A Dynamic Signal Analyzer on the modified EZ45-8G
I've added a secondary top to the cart, along with a
wood stop strip at the rear.
I also had to relocate the normal stop bar.
EZ45-6G cart with an Advantest R3463 spectrum analyzer on
top, and an Elecraft K2 on the bottom shelf, along with a Bird 100 watt 30 dB
The EZ45-6G is 15" wide x 19" deep and
fits most instruments.
You can see how the stop bar works with a typical size
instrument. The bar rotates up and the rubber pad contacts the instrument's
25 July 2007
Kees, K5BCQ, loaned me a Triad MIL-T-27E-SP68 audio
transformer to measure, as a possible audio isolation transformer to use with a
Softrock receiver. The transformer has, when properly terminated, a good
frequency response, up to 100 KHz, but you must ensure it is terminated
with the 10K design impedance to avoid a nasty high frequency resonance. Some
sound cards will present the required 10KΩ impedance to the transformer, whilst
others, such as the E-MU 0202 unit I use, would require auxiliary terminating
resistors, as the card's input impedance is 1 MΩ.
I've added the measured data, along with an LTSPICE model
of the MIL-T-27E-SP68, to my Softrock page.
Use the bookmarks at the top of the page to jump directly to the data.
23 July 2007
I've added details to the
Softrock page on a broadband 1.8 MHz - 30 MHz 1:1 transformer I use to to
isolate the antenna from my Softrock receiver.
The design may be of use for other purposes, of course.
22 July 2007
I've added a frequency response plot to my
Softrock page of the Triad 5-58X 600Ω : 600Ω
audio transformer that I use to isolate the receiver's I and Q channel outputs.
Unfortunately, these transformers seem to no longer be available, but they have
20 July 2007
I received three diamond core drills this morning and
tried them as PCB pad cutters. They work well, and better than my home made pad
cutter. I'll add more discussion at the Prototyping
Pads cut with 5 mm (3/16"), 7 mm (1/4") and 8 mm (5/16")
diamond core drills.
The amount of copper removed
with the two 1/4" drills differs slightly with the depth of cut. It's
important not to cut too deeply as there's a risk of removing all the copper
if the disk is snagged by the drill.
18 July 2007
The last few days have been devoted to working with 21.4
MHz monolithic crystal filters from ECS, ECS-21K7.5A and ECS-21K7.5B, in
for the data sheet. The 'A part is a two-pole filter in a single UM-1 case
(similar to an HC-49/U crystal case, but smaller)and the 'B part is two matched
two-pole filter modules, each in an UM-1 case. Both have a 7.5 KHz 3 dB
bandwidth, although different skirt response, as one might imagine.
The data sheet would have you believe that these filters
can be cascaded with a simple mesh coupling capacitor to ground between each
pair of 2-pole section. Don't believe it. Or, at least I couldn't make it work
and a telephone conversation with an applications engineer at ECS lead me to
believe it was not possible either. Rather, some isolation between filter
sections is required. In the case of the 'B filters, each pair of 2-pole
elements requires isolation. I believe isolation is also required between each
2-pole element if one is using 'A 2-pole sections as well.
Ideally, the isolation is provided by an amplifier with
high IP3 and low gain. Only four or five dB is required to offset an 8-pole
filter loss, which is about 2 to 2.5 dB per 4-pole group. For testing purposes,
I used an 850 ohm, 6 dB resistive pad which provided enough isolation to do the
I also tried L-networks, pi-networks and transformers to
match the filter impedance with my 50 ohm test gear. Although all worked, the
transformers worked the best. To match 50 ohms to 850 ohms for the 'B filters,
for example, I used an FT37-61 core with 6 turns on the 50 ohm side and 25 turns
on the 850 ohm side, for a theoretical 50 ohm : 868 ohm ratio. Plenty close
Following the data sheet method, without isolation between
filter sections, produced poor skirt rejection and a lumpy passband, as shown
below. I could make the flank spur move closer or further from the center, but
it would never move where it belonged, merged into the central passband.
With a resistive pad to isolate the two 4-pole filter sections, the response
looks like it should�flat top and smooth flanks, as reflected below.
Note the price paid for the resistive pad. The insertion
loss is now 11.5 dB, whilst in the earlier direct-connection design, the loss is
about 4 dB. Hence, an isolation amplifier has a certain appeal as it would
remove the excess loss whilst providing even better section-to-section
isolation. This layout also has excessive filter blow-by which is probably
related to the construction, most likely ground loop coupling.
The top sweep shows the flanks as being noise limited,
while the lower sweep shows clear evidence of blow-by.
Prototype filter with resistive pad isolation. Construction
15 July 2007
Perhaps the most common trait of ham operators is that of
a pack rat. It's hard to throw away something that may be used some day. When
searching in the garage today for a hose nozzle, I ran across a coffee can with
a couple hundred 1/2 watt International Resistance Co., carbon composition
resistors. These resistors were new around 1960, as I recall acquiring them not
long after that date, and are unused. Hence, they should not have drifted due to
conditions inside equipment.
I measured a selection of these resistors today and found
that unlike wine, carbon composition resistors do not get better with age. More
details at Carbon Composition
14 July 2007
I've added a section to my
Prototyping page discussing the "island cutting" alternative to Manhattan
construction. It works well, and avoids the perennial problem of breaking the
super glue bond while soldering to a Manhattan pad. The main disadvantage is
that you have to plan the construction and cut all the islands at once, or at
least that's the safest approach. Now, it may well be that this is not a
disadvantage at all, but rather enforces proper planning. I'll leave that call
21.4 MHz crystal filter board built with island cutting
12 July 2007 1400 EDT
If you own HP test equipment, it may have a NiCd battery
requiring replacement. Before nonvolatile EEPROM memory became available, HP
(and others) used rechargeable NiCad batteries to back up instrument settings
held in ordinary RAM. K8AQC loaned me his HP8116A digital function generator a
couple weeks ago and I noticed it failed to recall the last parameter settings,
and when powered on would default to 1 KHz sine.
Opening the 8116A's top cover showed why it was not recalling the last
parameters. This particular instrument was manufactured in 1986, and had the
original 1986 NiCad battery, which had corroded and leaked from both ends.
Fortunately, the corrosion was confined to the immediate area and had not
damaged adjacent traces or ICs.
Both ends of the battery leaked and corroded the top solder
Although I'm sure original equipment NiCad batteries are available, neither
Mouser nor DigiKey stocked them. Rather than search, I made up an
exact replacement from two Sanyo N-110AA NiCad cells obtained from Mouser. These
are 110 mAh NiCad cells, identical in physical size to the two cells used in the
original GE battery.
The original GE NiCad next to the two Sanyo replacement
To make the replacement battery, I soldered the two Sanyo cells in series and
encapsulated them with a short length of white heat-shrink tubing.
Replacement battery in place. The rectangular pads on the
PCB's top surface have some residual corrosion damage. Fortunately, the bottom
pads, where the tabs are soldered, were clean and no damage to the PCB was
12 July 2007 0745 EDT
I've measured the Softrock Lite 6.2 10 MHz band receiver's
input impedance and VSWR and posted the results at my
Softrock Lite 6.2 page.
11 July 2007 PM
I've tried two software defined radio programs from I2PHD
today, Winrad and SDRadio. Comments at my
Softrock Lite 6.2 page.
I've also added bookmarks and jumps on the Softrock page
for easier navigation.
11 July 2007
I've added a few paragraphs and photos to my
prototyping page showing a BNC mounting bracket I
designed and made today. The brackets were spurred when I started tinkering with
RF breadboarding a "Mode H" switched mixer and found the small PCB holders I
normally used too small for all the circuitry destined to be part of the
The "mode H" or switched mixer was developed by Colin
Horrabin, G3SBI, and offers extremely high third-order intermodulation intercept
figures. Readers interested in learning more about Mode H mixers might find
Martein's (PA3AKE) page
http://www.xs4all.nl/~martein/pa3ake/hmode/index.html a useful reference. As
I have some actual results, I'll provide my measurements as well. I do not
intend, however, to do the exhaustive optimization Martein has performed.
The bracket is for larger experimental layouts than are
accommodated in the small holders I've made. It's inconvenient to get RF signals
or audio signals in and out of a larger breadboard without making permanent
cable attachments, so my thought was a small aluminum bracket to hold a
single-hole mount BNC connector. The bracket can be attached to the edge of the
PCB breadboard for a mechanically sound connection. I made 10 of these in about
five hours today with the milling machine and lathe. It's not the highest
quality machine work I've ever done, but these are more than adequate for the
BNC mounting bracket for Manhattan-style breadboarding on a
scrap piece of board stock.
The intended use is for larger Manhattan-style layouts.
A variant design would mount the BNC vertically, so that it might be dropped
into the middle of Manhattan board, rather than at the edges as these
10 July 2007
If your oscilloscope is to work to the limits of its
specifications, it's important to use the proper probe. For most work, a 10X
probe, with a frequency specification matched to your oscilloscope is suitable.
Tektronix has an excellent tutorial on the care and feeding of oscilloscope
probes at www.tek.com. Use the search tool at
Tektronix's home page and search for "ABC of Probes." You may have to provide
some registration information before obtaining access to the tutorial.
The probe tip itself is only half the equation, however,
as the ground connection is also critical for the best results. I was looking at
a 20 MHz logic level clock oscillator today and thought it would make a useful
illustration of scope probe grounding. The oscilloscope is a Tektronix 2246, 100
MHz analog, with Tektronix P6106 10x probes.
Oscillator output with the normal long ground lead clipped
on the oscillator ground pin.
oscillator should have a flat top and bottom square wave with 5 V PP, we see a
spiked waveform over 7.5 V PP.
The oscilloscope image above shows signs of major ground lead ringing or
oscillation. The fast 20 MHz clock's fast rise/fall is shock exciting the
resonant circuit consisting of the inductance in the 6" ground lead and stray
capacitance. This is also known as "ground bounce."
We can reduce the ground lead inductance in the normal fashion, by shortening
it. And, I don't mean using a 4" ground lead instead of a 6" lead. Instead, slip
the hook sleeve off your scope probe. You should see a sharp pointed pin for the
center lead with a surrounding metal shell exposed. The metal shell is the probe
ground. (Some older probes do not have a removable sleeve. I have a couple of HP
probes from the 1960's of one piece construction, but probes manufactured in the
last 20 or 25 years are almost always of removable sleeve design, even
inexpensive generic probes.)
Your probe kit may have come with a wire coil or spring
type device, shown below slipped over the probe. This is a suitable short ground
connection for medium speed circuits. (There are even shorter probe arrangements
for faster transitions.)
Tektronix P6106 probe with sleeve removed and grounding
clip in place.
If your probe set did not include a ground adapter of this type, you can easily
make one from a short length of bare wire.
Home made spring ground adapter. I've soldered one end to
the oscillator's ground pin. (It goes without saying, of course, don't solder
to your scope probe!)
Clock output with the home made spring grounding adapter.
Although the on-screen readout says 5.87 V PP, counting
graticule lines shows it to be closer to 5.0 V.
The improvement should be obvious. There's still a bit of ringing and tilt, but
the waveform now looks like a square wave should.
you are looking at an audio signal, or one at a couple MHz or less, a long
ground lead won't usually be a problem. However, at higher frequencies and to
see the best edges on a waveform with fast rise/fall times, you have to pay
careful attention to your probe's ground connection.
One final reminder�even a 10X probe has a low input
impedance at high frequencies, as it is shunted by a few pF capacitance. By the
time you reach 30 or 40 MHz, even a few pF shunt capacitance represents a few
hundred ohm impedance. The shunt capacitance isn't too important in looking at
the 20 MHz TTL clock oscillator, but it can cause problems in other
circumstances. Tektronix covers this issue, along with many others, in the "ABC
of Probes." If you have not yet read it, do so.
09 July 2007
I've completed third order intermodulation measurements on the Softrock Lite
6.2 receiver, as well as minimum discernable signal and low frequency noise
(around the 0 Hz IF). The data is at the
07 July 2007
I recently purchased a used General Radio GR 1658
Digibridge and have been experimenting with it today. The 1658 reads resistance,
capacitance and inductance with quite decent accuracy, 0.1% in many cases. The
test frequencies are 100 Hz and 1000 Hz with my machine (it is set for European
frequencies) so it's not really suitable for very small capacitance and
inductance values in the ranges we often use in RF work, but it's still quite
handy to have around the shop.
In July 2001, I measured the effect of bias voltage on
capacitance, some of which data is presented at my
Capacitor Voltage Change page. Today,
I've taken more data with the 1658 Digibridge and expanded the page to include
more capacitor types (electrolytic and tantalum) and some smaller value (1000 pF)
parts of the type we use in radio work, including polystyrene, mica,
C0G/P0 dielectrics. I've added the new data to the Capacitor Voltage Change
page, now about double its prior length.
I've also tried
KGKSDR software with my
Softrock 6.2 with poor results�way too much CPU resources required to be usable.
Details at the Softrock Lite 6.2 page.
06 July 2007 PM
It's been a busy day. The E-MU 0202 sound card arrived and
I've had a couple hours to work with it and the Softrock lite 6.2 receiver. I've
extensively modified (rewritten) the Softock
Lite 6.2 page to reflect what I've found today, but the bottom line is that
the E-MU 0202 provides better performance in several respects, including a
maximum 192 kb/s sample speed. However, the added CPU resources required to
process the higher sample rate, and the associated graphical activity, really
tax my SX260's CPU resources, upwards of 85% CPU usage under certain
06 July 2007 AM
I thought I had damaged my 10 MHz Softrock lite v. 6.2
receiver by overloading the signal input with a +20 dBm signal during testing,
so this morning I went through it with an oscilloscope, voltmeter and signal
generator (at -10 dBm this time). I found some odd signals and voltage
readings centered around U4 (an FST3253 fast bus switch used as the mixer) and
U5, a TLV2462 dual op-amp active filter. When building the board, I must have
made a poor solder joint at U5, pin 3 or 5. Reheating both pins fixed the
problem and the receiver seems to be working properly.
Later this morning, I should receive the E-MU 0202
USB 2.0 external sound card to use with the two Softrock receiver boards.
Before trouble shooting the receiver board, I made
up a ground clip to header pin adapter. I acquired several of these along with
some Tektronix scope probes, but I thought it would be a useful quick project to
document on these pages.
The device I modeled it after is a "SAFTLEED" as shown
below. The black projection is a receptacle for a standard 0.025" square header
pin. The white object is an insulated wire connector that serves as a insulating
shell to protect the wire loop. The purpose of this gadget is to provide a safe
connection for an oscilloscope ground clip to a standard header pin. Most
oscilloscope ground leads have an alligator clip that's too large to easily
connect to a header pin without risking shorting adjacent pins.
To use the adapter, you slide the receptacle over the
header pin and clip the probe's alligator ground clip onto the wire loop inside
the shell. It works well, and can also be used for signal connections, as it's
often more convenient than trying to clip the scope probe tip onto a header pin.
I have one in red, one green and one dark blue, which makes distinguishing
You can read the details in making one of the adapters,
and see how it is used, by clicking here or
going to the link Header Adapter at the top left of this page.
Original "SAFTLEED" as supplied with Tektronix probe set.
My home brew version of the SAFTLEED.
04 July 2007 0800 EDT
I realized last evening that the 21.4 MHz filter
intermodulation data has a flaw--the two signal generators were spaced far
enough apart that the intermodulation product fell outside the filter passband.
Hence, an IMD product generated in the first filter section would be attenuated
by the second filter section.
This morning, I re-ran the data with the two signal
generators at 21.401 MHz and 21.399 MHz, i.e., 2 KHz spacing centered in the
filter passband. The 3rd order products will thus be at 21.397 and 21.403 MHz,
within the filter passband. We should, therefore, see IMD caused by any filter
I ran tests with filter inputs of +6 dBm, -4 dBm and -14
dBm, with the -4 and -14 dBm values obtained by inserting 10 and 20 dB
attenuators between the combiner output and filter input.
To show the increase in intermodulation products, I ran
the spectrum analyzer in dual trace mode. The reference trace for each level is
a 2 dB pad, which almost exactly duplicates the filter's 1.9 dB measured
insertion loss. The second trace is the filter. Using a 2 dB pad allows the
trace data to match and presents the spectrum analyzer with the same input level
as when the filter is in place.
Spectrum analyzer data is captured with a Prologix GPIB-to-USB
interface card, and rendered with KEFX's 7470 emulation program.
The blue trace is the saved 2 dB pad data, whilst the
black trace is the 21.4 MHz filter. Where the two traces overlay each other, the
blue will be on top.
The first test is with zero dB attenuation, which presents
the filter with an input signal of about +8 dBm. Note that not only are 3rd
order intermodulation products visible, so are 5th order products.
IP3 is: +23 dBm, reasonably close to the +26 of
yesterday's data, but worse, as expected since the IMD product is no longer
masked by filter attenuation. (IP3 based on filter output; add 2 dB if an input
reference is desired.)
The IP3 of the test setup is +42 dBm, with the limiting
factor being combiner isolation and mixing in the signal generators.
With 10 dB reduction (to -2 dBm) to the filter's
input. Note that the 5th order products theoretically will drop 50 dB with 10 dB
change in input level, which places them below the noise level in this
particular test setup.
The filter's IP3 is +24 dBm.
The final plot is with 20 dB attenuation, so the filter input is -12 dBm. This
is slightly below the filter manufacturer's -10 dBm recommended maximum input
The IP3 is +22 dBm.
Variations of ±1 dB can be disregarded for a quick test
such as this. Hence, the filter shows an essentially constant IP3 of about
+23 dBm for input levels from -12 dBm to +8 dBm. The filter's IP3 values are
well behaved and do not show significant changes in computed IP3 with different
drive at the levels tested.
+23 dBm is not a bad IP3 level, but if to be used in a
receiver with a target IP3 of, for example, +30 dBm, this filter would be the
03 July 2007 1800 EDT
In my 21.4 MHz crystal filter discussion earlier today, I
mentioned that monolithic crystal filters are prone to intermodulaton
distortion, particularly when driven hard, i.e., above -10 dBm. I've made
a quick IP3 measurement on the crystal filter assembly this evening.
The equipment is:
- Signal generator 1: HP 8640B
- Signal generator 2: Panasonic VP8191A
- 6 dB pads on both generator outputs Minicircuits CAT-6
- Combiner: Minicircuits LRPS-2-1A
- Spectrum analyzer: Advantest R3463
Both signal generators were set to +18 dBm, one slightly
bellow the filter center and the other slightly above. The net loss through the
6 dB pads and the LRPS-2-1A combiner is a bit over 9 dB.
The spectrum analyzer image below shows the IMD floor of
the test setup. The filter is replaced with a BNC barrel connector in this test.
The IP3 is approximately +43 dBm.
The figure below replaces the BNC barrel with the 21.4 MHz
crystal filter. Note the significant increase in the two IMD spurious signals
generated by the filter.
The IP3 is approximately
+26 dBm, 17 dB worse than the measurement floor.
An IP3 of +26 dBm is probably acceptable for the application.
For one thing, the the test drives the filter input at around +9 dBm, which is
well above the level that will be applied in practice. I'll run further
tests as I approach the final filter configuration, and see how non-linear the
IMD products are with changes in drive level. A cubic law device has a 1:3
change ratio, i.e., 1 dB increase in input signal causes a 3 dB increase
in 3rd order intermodulation products. I suspect the crystal filter will not
follow a perfect cubic law and that with an input level more realistic, say -10
to -20 dBm, the IMD performance will be much better.
03 July 2007
I've taken a break from programming for the day, instead
tinkering with a 21.4 MHz crystal filter. The next generation panadapter I'm
working towards will use a 21.4 MHz 1st IF, with a relatively broad filter, 7.5
KHz or so, and a 2nd IF at 12 KHz with DSP. I'm uncertain whether to use a full
I&Q 2nd IF or go with a conventional design and won't make that decision until
ECS has an attractive series of
21.4 MHz crystal filters. The filters are individual two-pole monolithic
quartz units that may be cascaded in 2, 4, 6, 8 or 10 pole configurations. I
built up a 4-pole 7.5 KHz bandwidth filter today, and measured several key
These filters are intended for AM or FM communications (the 7.5 KHz bandwidth
filters are likely intended for VHF-AM, such as air-to-ground communications)
and as such have a flat top response. That's not what one wants for a spectrum
analyzer, but the 21.4 MHz filter will serve as a roofing filter for the DSP,
which will have a traditional Gaussian response. With wider DSP bandwidths, the
response will be compromised, but still usable. I believe a 3 to 5 KHz Gaussian
bandwidth will look good, but the maximum resolution bandwidth of, say, 7.5 KHz,
will be more flat topped than rounded.
My concern with these filters has been ultimate rejection and flank
bandwidth. With a 12 KHz second IF and without I&Q demodulation, image response
can be a problem, particularly with wider bandwidths. I would like to see 80 to
90 dB at ±16 KHz from center. That may
be achievable, based on today's data, with a 6 or 8 pole filter.
A problem with monolithic filters is intermodulation, due to
the crystal's parameters varying with drive, the so-called "drive level
dependency." I will measure the filter's IMD tomorrow, although that may be
The three trimmers adjust the filter's input, output and inter-stage coupling.
Tweaking these is necessary to obtain the desired passband response. The two
toroids are part of an L-network to match the filter's 850 ohm impedance to 50
Four-pole monolithic crystal filter. Each "can" has a
two-pole filter. The second is mostly obscured by the shield but can be seen
in part at the center of the shield.
is the one described at my prototyping page. I
have a box of close to three dozen small circuits for this fixture, built over
the last 10 years.
ECS-21K7.5 filter passband. Measured insertion loss 1.9 dB
(spec is 2.5 dB with 1 dB ripple). I adjusted the trimmers for maximum flatness,
which, I suspect, degraded the flank selectivity and slightly broadened the
passband, at least partially shifting the filter from a Chebyshev to a
Butterworth response. This is, by the way, the easiest crystal filter I've ever
built, as ECS has done all the hard work.
is taken with an HP8752B vector network analyzer, time base locked to an HP
5065A rubidium frequency standard.
Group delay. Not too bad, about 75 μs variation in the passband.
Expanded span view. The flanks are symmetrical and show about 70 dB rejection at
this scale. A wider span view shows ultimate rejection in the 90 dB range, which
is the limit of the test configuration.
Over a 5 MHz span no signs of spurious responses. It will likely be useful to
back up the 21.4 MHz crystal filter with an LC bandpass filter to avoid overtone
response. To some extent, the input and output LC matching network provides low
pass filtering, but additional LC shaping will be useful.
03 July 2007
Further research explains why Rocky consumes
disproportionate CPU resources running on my Dell SX260. Although the SX260 has
an Intel graphic controller on the mother board (the SX260 is a zero slot
machine, so no substitute graphics card is possible), it does not have dedicated
graphics memory. Rather, 32 MB of the main memory is allocated to the graphic
controller. This cost savings measure extracts a major price on graphics speed.
A graphic intensive program, such as Rocky, runs much slower than on a PC with a
normal graphics card.
I've also ordered an external sound card with USB 2.0
interface to see how much of the noise and zero-Hz artifacts are related to the
SX260's internal SoundMax sound card.
02 July 2007
Rocky v. 3.3.2
is now available. It fixes the S-meter problem. CPU resources are still an issue
with my installation, and seem to be related to screen size. With the smallest
reasonable screen, CPU requirements are 24%. That's high, but better than 53%.
With Rocky minimized to invisibility, CPU resources are 15%. I've added
intermediate data points to my Softrock page.
I've also run a quick test with an antenna isolation transformer and battery
power for my Softrock receiver, without noticeable improvement.
More recent information ... Rocky loaded on my Gateway
laptop is running around 10% or less CPU resources, even running full screen. My
SX260 must be severely graphics-limited as the difference between the two
computers is far more than might be attributable to CPU speeds or cores.
02 July 2007
I've installed Rocky, v. 3.3.1, for Softrock
receiver, and update the Softrock page to
reflect my initial impressions of 3.3.1. Version 3.3.1 is a bug fix release, but
the problems I observed are not addressed with v. 3.3.1.
01 July 2007 PM
I've received feedback on my Softrock page and have made
changes to reflect those comments, and also my trial of PowerSDR software as
well as Rocky. I like Rocky's user interface, but PowerSDR is not as demanding
on CPU resources and also is (so far, at least) free of stuttering and crashing,
I'm now able to make an HP8116A function generator and an
HP3456A voltmeter co-exist on the same GPIB bus. The data plot below of a 15 KHz
DSP-based filter, is taken with that combination.
The responses at 5 KHz and 7.5 KHz are from the 8116A's
third and second harmonics, respectively. Since the 3456A voltmeter is a
broadband detector, it responds to any signals that make it through the 15 KHz
filter. In this case, when set at 5 KHz, the 8116A's third harmonic is about -50
dBc, which is more than strong enough to be read. Even the second harmonic
(8116A generator at 7.5 KHz) can be seen, and it's about -55 dBc. When measuring
filter response with a broadband detector, you must keep generator harmonics in
The June 2007 Updates page is now at the June 2007 Archive
and may be read by clicking here, or via the Archive
links at the top of this page. It's now one year since I started the Clifton
Laboratories web page.
The last few days have been busy with two activities.
First, one architecture I'm considering for the DSP-based
panadapter is an I&Q down-converting design. Conventional swept spectrum
analyzer front end, but the second IF selectivity and log conversion via DSP.
This has some similarity to the Softrock receiver, so I purchased
Softrock 6.2 Lite kits for 7 MHz and
10 MHz bands and built them over the last couple days. I've added a page with
photos and some measurements, linked here or
via the main navigation menu at the top left of this page.
I got a bit carried away making preliminary measurements
on the 10 MHz receiver and blew the FST3253 bus switch mixer by over driving it
with an HP 8640B signal generator. But, I've been listening to a 40 KHz swath of
the 7 MHz band with the second kit since yesterday. I'm using
VE3NEA's Rocky software and a Dell
Opti-Plex SX260's built-in soundcard. It's not the quietest soundcard and the
Softrock receiver board is laying on my programming table, but I can still hear
a signal at 0.3 uV. Although the SX260 has 2 GB RAM and a 2.3 GHz processor,
Rocky consumes about 50% of CPU clock cycles. Rocky is doing a lot of processing
so it's understandable that it grabs quite a bit of CPU time. However, it makes
other programming a bit sluggish at times.
The second project involves making a friend's HP8116A
function generator play with a
Prologix USB-to-GPIB interface card. Although the 8116A will accept message
and properly set the instrument's parameters, reading back settings has been
difficult, as the response strings are embedded in a sea of "NO MESSAGE"
messages emitted by the 8116A. I've discussed the problem with Prologix's
developer and a new firmware release (version 4.65) for the USB-GPIB adapter has
solved the problem, or at least made it possible to easily read the 8116A's
responses, while filtering out the NO MESSAGE fillers.
If you have test gear with GPIB communications, you should
consider Prologix's adapter. It's reasonably priced and the times I've needed
support, it has been provided very quickly and my questions/problems resolved in